GPS - đường dẫn quán tính và hội nhập P4

Global Positioning Systems, Inertial Navigation, and Integration, Mohinder S. Grewal, Lawrence R. Weill, Angus P. Andrews Copyright # 2001John Wiley & Sons, Inc. Print ISBN 0-471-35032-X Electronic ISBN 0-471-20071-9 4 Receiver and Antenna Design 4.1 RECEIVER ARCHITECTURE Although there are many variations in GPS receiver design, all receivers must perform certain basic functions. We will now discuss these functions in detail, each of which appears as a block in the diagram of the generic receiver shown in Fig. 4.1. 4.1.1 Radio-Frequency Stages (Front End) The purpose of the receiver front end is to ®lter and amplify the incoming GPS signal. As was pointed out earlier, the GPS signal power available at the receiver antenna output terminals is extremely small and can easily be masked by inter-ference from more powerful signals adjacent to the GPS passband. To make the signal usable for digital processing at a later stage, RF ampli®cation in the receiver front end provides as much as 35±55dB of gain. Usually the front end will also contain passband ®lters to reduce out-of-band interference without degradation of the GPS signal waveform. The nominal bandwith of both the L1 and L2 GPS signals is 20MHz (10MHz on each side of the carrier), and sharp cutoff bandpass ®lters are required for out-of-band signal rejection. However, the small ratio of passband width to carrier frequency makes the design of such ®lters infeasible. Consequently, ®lters with wider skirts are commonly used as a ®rst stage of ®ltering, which also helps to prevent front-end overloading by strong interference, and the sharp cutoff ®lters are used later after downconversion to intermediate frequencies (IFs). 80 4.1 RECEIVER ARCHITECTURE 81 Antenna RF stage First IF stage Second IF stage First Mixer BPF Amp BPF Second Mixer Amp BPF Amp A/D converter LO LO Reference oscillator Frequency synthesizer Digitized IF signal Clocks Interrupts Code aquisition/tracking Carrier acquisition/tracking Message bit synchronization Navigation message demodulation Code/carrier pseudoranging Delta-range measurements H/W & S/W Signal Proccessing External inputs (INS, altimeter, Loran-C, clock aiding) Navigation outputs (position, velocity, time, fault, detection/isolation) Fig. 4.1 Generic GPS receiver. 4.1.2Frequency Downconversion and IF Ampli®cation After ampli®cation in the receiver front end, the GPS signal is converted to a lower frequency called an intermediate frequency for further ampli®cation and ®ltering. Downconversion accomplishes several objectives: 1. The total amount of signal ampli®cation needed by the receiver exceeds the amount that can be performed in the receiver front end at the GPS carrier frequency. Excessive ampli®cation can result in parasitic feedback oscillation, which is dif®cult to control. In addition, since sharp cutoff ®lters with a GPS signal bandwidth are not feasible at the L-band, excessive front-end gain makes the end-stage ampli®ers vulnerable to overloading by strong nearby out-of-band signals. By providing additional ampli®cation at an IF different from the received signal frequency, a large amount of gain can be realized without the tendency toward oscillation. 2. By converting the signal to a lower frequency, the signal bandwidth is unaffected, and the increased ratio of bandwidth to center frequency permits the design of sharp-cutoff bandpass ®lters. These ®lters can be placed ahead of the IF ampli®ers to prevent saturation by strong out-of-band signals. The ®ltering is often by means of surface acoustic wave (SAW) devices. 82 RECEIVER AND ANTENNA DESIGN 3. Conversion of the signal to a lower frequency makes the sampling of the signal required for digital processing much more feasible. Downconversion is accomplished by multiplying the GPS signal by a sinusoid called the local oscillator signal in a device called a mixer. The local oscillator frequency is either larger or smaller than the GPS carrier frequency by an amount equal to the IF. In either case the IF signal is the difference between the signal and local oscillator frequencies. Sum frequency components are also produced, but these are eliminated by a simple band-pass ®lter following the mixer. An incoming signal either above or below the local oscillator frequency by an amount equal to the IF will produce an IF signal, but only one of the two signals is desired. The other signal, called the image, can be eliminated by bandpass ®ltering of the desired signal prior to downconversion. However, since the frequency separation of the desired and image signals is twice the IF, the ®ltering becomes dif®cult if a single down-conversion to a low IF is attempted. For this reason downconversion is often accomplished in more than one stage, with a relatively high ®rst IF (30±100MHz) to permit image rejection. Whether it is single stage or multistage, downconversion typically provides a ®nal IF that is low enough to be digitally sampled at feasible sampling rates without frequency aliasing. In low-cost receivers typical ®nal IFs range from 4 to 20MHz with bandwidths that have been ®ltered down to several MHz. This permits a relatively low digital sampling rate and at the same time keeps the lower edge of the signal spectrum well above 0Hz to prevent spectral foldover. However, for adequate image rejection either multistage downconversion or a special single-stage image rejection mixer is required. In more advanced receivers there is a trend toward single conversion to a signal at a relatively high IF (30±100MHz), because advances in technology permit sampling and digitizing even at these high frequencies. Signal-to-Noise Ratio An important aspect of receiver design is the calculation of signal quality as measured by the signal-to-noise ratio (SNR) in the receiver IF bandwith. Typical IF bandwidths range from about 2MHz in low-cost receivers to the full GPS signal bandwidth of 20MHz in high-end units, and the dominant type of noise is the thermal noise in the ®rst RF ampli®er stage of the receiver front end (or the antenna preampli®er if it is used). The noise power in this bandwidth is given by N kTeB 4:1 where k 1:3806 1023 J=K, B is the bandwidth in Hz, and Te is the effective noise temperature in degrees Kelvin. The effective noise temperature is a function of sky noise, antenna noise temperature, line losses, receiver noise temperature, and ambient temperature. A typical effective noise temperature for a GPS receiver is 513K, resulting in a noise power of about 138:5dBW in a 2-MHz bandwidth and 128:5dBW in a 20-MHz bandwidth. The SNR is de®ned as the ratio of signal power to noise power in the IF bandwidth, or the difference of these powers when 4.1 RECEIVER ARCHITECTURE 83 expressed in decibels. Using 154:6dBW for the received signal power obtained in Section 3.3, the SNR in a 20-MHz bandwidth is seen to be 154:6 128:5 26:1dB. Although the GPS signal has a 20-MHz bandwidth, about 90% of the C=A-code power lies in a 2-MHz bandwith, so there is only about 0.5dB loss in signal power. Consequently the SNR in a 2-MHz bandwidth is 154:6 0:5 138:5 16:6dB. In either case it is evident that the signal is completely masked by noise. Further processing to elevate the signal above the noise will be discussed subsequently. 4.1.3 Digitization In modern GPS receivers digital signal processing is used to track the GPS signal, make pseudorange and Doppler measurements, and demodulate the 50-bps data stream. For this purpose the signal is sampled and digitized by an analog-to-digital converter (ADC). In most receivers the ®nal IF signal is sampled, but in some the ®nal IF signal is converted down to an analog baseband signal prior to sampling. The sampling rate must be chosen so that there is no spectral aliasing of the sampled signal; this generally will be several times the ®nal IF bandwidth (2±20MHz). Most low-cost receivers use 1-bit quantization of the digitized samples, which not only is avery low cost method of analog-to-digital conversion, but has the additional advantage that its performance is insensitive to changes in voltage levels. Thus, the receiver needs no automatic gain control (AGC). At ®rst glance it would appear that 1-bit quantization would introduce severe signal distortion. However, the noise, which is Gaussian and typically much larger than the signal at this stage, introduces a dithering effect that, when statistically averaged, results in an essentially linear signal component. One-bit quantization does introduce some loss in SNR, typically about 2dB, but in low-cost receivers this is an acceptable trade-off. A major disadvantage of 1-bit quantization is that it exhibits a capture effect in the presence of strong interfering signals and is therefore quite susceptible to jamming. Typical high-end receivers use anywhere from 1.5-bit (three-level) to 3-bit (eight-level) sample quantization. Three-bit quantization essentially eliminates the SNR degradation found in 1-bit quantization and materially improves performance in the presence of jamming signals. However, to gain the advantages of multibit quantiza-tion, the ADC input signal level must exactly match the ADC dynamic range. Thus the receiver must have AGC to keep the ADC input level constant. Some military receivers use even more than 3-bit quantization to extend the dynamic range so that jamming signals are less likely to saturate the ADC. 4.1.4 Baseband Signal Processing Baseband signal processing refers to a collection of high-speed real-time algorithms implemented in dedicated hardware and controlled by software that acquire and track the GPS signal, extract the 50-bps navigation data, and provide measurements of code and carrier pseudoranges and Doppler. 84 RECEIVER AND ANTENNA DESIGN Carrier Tracking Tracking of the carrier phase and frequency is accomplished by using feedback control of a numerically controlled oscillator (NCO) to frequency shift the signal to precisely zero frequency and phase. Because the shift to zero frequency results in spectral foldover of the signal sidebands, both in-phase (I) and a quadrature (Q) baseband signal components are formed in order to prevent signal information loss. The I component is generated by multiplying the digitized IF by the NCO output and the Q component is formed by ®rst introducing a 90 phase lag in the NCO output before multiplication. Feedback is accomplished by using the measured baseband phase to control the NCO so that this phase is driven toward zero. When this occurs, signal power is entirely in the I component, and the Q component contains only noise. However, both components are necessary both in order to measure the phase error for feedback and to provide full signal information during acquisition when phase lock has not yet been achieved. The baseband phase ybaseband is de®ned by ybaseband atan2I;Q 4:2 where atan2 is the four-quadrant arctangent function. The phase needed for feedback is recovered from I and Q after despreading of the signal. When phase lock has been achieved, the output of the NCO will match the incoming IF signal in both frequency and phase but will generally have much less noise due to low-pass ®ltering used in the feedback loop. Comparing the NCO phase to a reference derived from the receiver reference oscillator provides the phase measurements needed for carrier phase pseudoranging. Additionally, the cycles of the NCO output can be accumu-lated to provide the raw data for Doppler, delta-range, and integrated Doppler measurements. Code Tracking and Signal Spectral Despreading The digitized IF signal, which has a wide bandwidth due to the C=A- (or P-) code modulation, is completely obscured by noise. The signal power is raised above the noise power by despreading, in which the digitized IF signal is multiplied by a receiver-generated replica of the code precisely time aligned with the code on the received signal. Typically the individual baseband I and Q signals from the controlled NCO mixer are despread in parallel, as previously shown in Fig. 3.13. The despreading process removes the code from the signal, thus concentrating the full signal power into the approximately 50-Hz baseband bandwidth of the data modulation. Subsequent ®ltering (usually in the form of integration) can now be employed to dramatically raise the SNR to values permitting observation and measurement of the signal. As an example, recall that in a GPS receiver a typical SNR in a 2-MHz IF bandwidth is 16:6dB. After despreading and 50-Hz low-pass ®ltering the total signal power is still about the same, but the bandwidth of the noise has been reduced from 2MHz to about 50Hz, which increases the SNR by the ratio 2 106=50, or 46dB. The resulting SNR is therefore 16:6 46:0 29:4dB. ... - tailieumienphi.vn